1. Field of the Invention
The present invention relates to orthogonal frequency division multiplexing (OFDM) communication systems and methods and to transmitters and transmitting methods for use in such systems.
2. Description of the Related Art
An OFDM system is a type of multi-carrier transmission system, in which a single data stream is modulated onto N parallel sub-carriers, each sub-carrier signal having its own centre frequency. The sub-carriers are transmitted together as a single combined OFDM signal, the individual sub-carrier signals being recoverable in an OFDM receiver due to the orthogonal property of the sub-carriers. Typically, the number N of sub-carriers may be relatively large, for example N=512.
OFDM systems have many advantages. Orthogonal frequency division multiplexing (OFDM) is a modulation technique that is based on traditional frequency division multiplexing (FDM), but is much more spectrally efficient than traditional FDM because the sub-channels (sub-carriers) may be spaced,much closer together in frequency (i.e. until they are actually overlapping) than in FDM systems (in which guard bands are required between each sub-channel).
The spectral efficiency of an OFDM system is achieved because the frequency of the respective sub-carriers are chosen such that they are mutually orthogonal (which means that they are mutually perpendicular in a mathematical sense), thereby allowing the spectrum of each sub-carrier to overlap the spectra of other sub-carriers without interfering with them. The effect of this mutually orthogonal nature of the OFDM sub-carriers is that the required bandwidth is greatly reduced, as compared to traditional FDM systems, by removing the guard bands and allowing the spectra of the sub-carriers to overlap one another.
Another advantage of OFDM is its resilience to multipath, which is the effect of multiple reflected signals hitting the receiver. OFDM systems are capable of overcoming the potential problems associated with multipath, such as interference and frequency-selective fading, because the total signal bandwidth (i.e. the amount of data to be sent in a given time interval) is divided over the plurality of sub-carriers. As a result of this parallel transmission, the duration of each OFDM data symbol may be relatively long, and each individual sub-carrier may have a relatively low information rate, thereby allowing the system to benefit from enhanced immunity to impulse noise and reflections.
OFDM's high spectral efficiency and resistance to multipath make it an extremely suitable technology to meet the demands of wireless data traffic. Today, the technology is used in such systems as asymmetric digital subscriber lines (ADSL) as well as in wireless systems such as IEEE 802.11a/g (Wi-Fi) and IEEE 802.16 (Wi-MAX). It is also used for wireless digital audio and video broadcasting. OFDM is currently one of the prime technologies being considered for use in future fourth generation (4G) networks.
In an OFDM communication system a plurality N of sub-carriers are employed to carry data from a transmitter to one or more receivers. The number N of sub-carriers may be relatively large, for example N=512. One problem which arises in OFDM communication systems is that a peak-to-average power ratio (hereinafter PAPR) tends to be high. The peak power increases generally according to the number of sub-carriers. For example, when the N signals carried respectively by the sub-carriers have the same phase, the maximum power of the resulting multi-carrier signal is increased to N times an average power. When the PAPR is high, an amplifier having a very wide dynamic range is required in the transmitter, which is undesirable.
Numerous techniques have been proposed to solve the problems with PAPR in OFDM communication systems. Before describing some of these techniques, the basic features of a transmitter for use in an OFDM communication system will be described with reference to FIG. 1.
FIG. 1 shows parts of an OFDM transmitter 10. The transmitter comprises a serial-to-parallel converter 12 which receives a serial stream of input symbols. Although not shown in FIG. 1, the input symbols may comprise data symbols, provided from one or more data sources, and pilot symbols provided from a pilot symbol generator (not shown). The serial-to-parallel converter divides the received input signals into groups of N input symbols and outputs the symbols X0 to XN-1 of each group in parallel. The transmitter 10 further comprises an IFFT processing unit 14 which subjects each group of N symbols X0 to XN-1 to N-point inverse fast Fourier transform (IFFT) processing to produce IFFT output signals x0 to xN-1 representing N digital samples of the time-domain signals. These IFFT output signals are subjected to parallel-to-serial conversion in a parallel-to-serial converter 16. After conversion into serial form, the IFFT output signals are applied to a cyclic prefix (CP) addition unit 18.
The CP addition unit 18 adds a cyclic prefix at the beginning of each OFDM symbol period (here the OFDM symbol period is N×Ts, where Ts, is the input symbol period (the period of each data symbol and each pilot symbol)). As is well known, in OFDM systems, orthogonality of sub-carriers is lost when multipath channels are involved. The addition of the cyclic prefix can restore the orthogonality at the receiver, although energy is wasted in the cyclic prefix samples.
The serial IFFT output signals with the added cyclic prefix are then applied to a windowing unit 20 which carries out a windowing operation for pulse shaping. The digital signal samples are then converted into an analog baseband signal in a digital-to-analog converter (DAC) 22 and the analog baseband signal is then supplied to a radio frequency (RF) unit 24. The RF unit 24 converts the baseband signal into a RF signal suitable for transmission to one or more receivers via an antenna 26 of the transmitter 10.
Next, phase adjustment techniques which have been proposed for solving the PAPR problem will be described with reference to FIGS. 2, 3 and 4.
FIG. 2 shows parts of an OFDM transmitter 40 adapted to carry out a partial transmit sequence (PTS) method.
In FIG. 2, a data source 42 supplies a serial stream of data symbols DS to a serial-to-parallel converter 46. The serial-to-parallel converter 46 also receives from time to time pilot symbols PS from a pilot symbol generator 44. The converter 46 combines the received data symbols DS and any pilot symbols PS and forms groups of input symbols. Each group is made up of N input symbols. The serial-to-parallel converter 46 outputs the N input symbols of each group in parallel to a symbol division unit 48. The symbol division unit 48 divides the N input signals of each group into M sub-groups (sometimes referred to as “sub-blocks”), each made up of L (=N/M) input symbols. Each of the M sub-blocks X1 to XM is subjected to L-point inverse discrete Fourier transform (IDFT) processing by a corresponding IDFT processing unit 501, to 50M. The IDFT output signals from each IDFT processing unit 50, which may also be referred to as “partial transmit sequences”, are supplied to a first input of a corresponding complex multiplier 521, to 52N. Each complex multiplier 521, to 52M also receives at a second input a phase adjustment factor b1, to bm. Each complex multiplier 521, to 52M adjusts the phases of all L IDFT output signals of the sub-block by the applied phase adjustment factor. The outputs of the complex multipliers 521, to 52M are then combined by a combiner 56.
The individual phase adjustment factors together form a phase vector b. An optimisation unit 54 stores a plurality of available phase vectors. The optimisation unit 54 receives the IDFT output signals from the IDFT processing units 501 to 50M and selects one phase vector from among the plurality of available phase vectors. The selected phase vector is the one that produces a sub-optimal phase-adjusted combination of the IDFT output signals of the different sub-blocks. This combination of N IDFT output signals, when transmitted, will have a desirably low PAPR.
The time-domain samples output from the combiner 56 are then subjected to parallel-to-serial conversion, cyclic prefix addition and the subsequent processing described previously with reference to FIG. 1 (units 16, 18, 20, 22 and 24 in FIG. 1).
The optimisation unit 54 also outputs identification information which may be transmitted to the receiver(s) as side information to enable the receiver(s) to identify the phase vector selected by the transmitter. The receiver can then carry out reverse processing to recover the data and pilot symbols.
In the PTS method, the amount PAPR reduction that is achieved depends on the number of sub-blocks. The greater the number of sub-blocks the larger the potential PAPR reduction that is achievable. However, increasing the number of sub-blocks increases the processing burden on the transmitter. For example, the complexity of the search space for the best combination of IDFT output signals increases exponentially as the number of sub-blocks increases, which is generally prohibitive.
Further information regarding PTS methods can be found in “A novel peak power reduction for OFDM”, S. H. Müller, J. B. Huber, PROC. IEEE PIRMC '97, Helsinki, Finland, September 1997, pp 1090-94, and in “A Comparison of peak power reduction reduction schemes for OFDM”, S. H. Müller, J. B. Huber, PROC. IEEE Globecom '97, Phoenix, Ariz., USA, pp 1-5.
Another phase adjustment technique which has been proposed is a selective mapping (SLM) method. FIG. 3 shows parts of an OFDM transmitter adapted to carry out an SLM method. In the transmitter 60 of FIG. 3, a serial stream of input symbols (data symbols and pilot symbols) is applied to a serial-to-parallel converter 62. The converter 62 outputs groups C of input symbols, there being N input symbols in each group C.
The transmitter 60 also comprises a phase vector storage unit 64 which stores a set of U available phase vectors P{tilde over (u)} (ũ=1, . . . U). These phase vectors are generated randomly and are statistically independent. Each phase vector is made up of N phase elements θ0, θ1, θ2, . . . θN-1,Pu=[ejφ0u, ejφ1u, . . . , ejφN-1u]  (1)assuming that φnuε(0,2π), u ε{1, . . . , U}.
Each phase element corresponds individually to one of the N sub-carriers and sets a phase adjustment to be applied by the transmitter to the corresponding sub-carrier for the group C of input symbols concerned.
The U phase vectors P1 to PU and the group C of input symbols are supplied to a candidate symbol generator 66. The candidate symbol generator calculates the vector product of the group C of input signals and each of the available phase vectors PU to produce U candidate OFDM symbols CS1 to CSU.
The candidate OFDM symbols CS1 to CSU are supplied to a selection unit 68. The selection unit 68 calculates a potential transmitted signal for each candidate symbol CS1 to CSU, according to the expression
                                                                                          s                  ⁡                                      (                    t                    )                                                  =                                                      1                                          N                                                        ⁢                                                            ∑                                              n                        =                        0                                                                    N                        -                        1                                                              ⁢                                                                  c                        n                                            ⁢                                              ⅇ                                                  j                          ⁢                                                                                                          ⁢                          2                          ⁢                                                                                                          ⁢                          π                          ⁢                                                                                                          ⁢                          Δ                          ⁢                                                                                                          ⁢                          ft                                                                                                                                ,                                                          0              ≤              t              ≤              T                                                          (        2        )            where C=(c0c1 . . . CN-1) represents a vector of N constellation symbols from a constellation. For the signal s(t) a measure of the PAPR is given by:
                    ξ        =                              max            ⁢                                                                            s                  ⁡                                      (                    t                    )                                                                              2                                            E            ⁢                          {                                                                                      s                    ⁡                                          (                      t                      )                                                                                        2                            }                                                          (        3        )            where E denotes expectation.
It will be appreciated that to calculate the potential transmitted signal for each candidate symbol an N-point IFFT operation is required. Accordingly, the selection unit 68 includes an IFFT processing unit 70 similar to the IFFT processing unit 14 described previously with reference to FIG. 1. The IFFT processing unit 70 needs to carry out U of the N-point IFFT operations and, optionally, the selection unit 70 may be provided with U IFFT processing units 701, to 70U to enable the operations to be carried out in parallel.
After calculating such a PAPR measure for each candidate symbol CS1 to CSU, the selection unit 68 selects the candidate symbol CSSLM which has the lowest PAPR measure. IFFT output signals xSLM corresponding to the selected candidate symbol CSSLM are output by the selection unit. These IFFT output signals xSLM are then subjected to parallel-to-serial conversion and CP addition, etc., as described previously with reference to FIG. 1.
Optionally, also, the selection unit 68 outputs the identity ũ of the selected phase vector, i.e. the phase vector corresponding to the selected candidate symbol CSSLM. This identity ũ may be transmitted as side information to the receiver(s) to enable the receiver(s) to carry out reverse phase adjustments corresponding to the phase adjustments applied by the transmitter using the selected phase vector Pu.
Although SLM offers a significant PAPR reduction, the size of the search space (the number of phase vectors which must be processed) is high to achieve PAPR reductions of the desired magnitude. This in turn implies that immense processing power is required in the transmitter and that the signalling overhead for transmitting the identity ũ may be undesirably high. Although it is possible to trade off additional receiver complexity with a reduction in the signalling overhead, for example by adopting a blind or semi-blind receiver, SLM methods involving high numbers of phase vectors are still complex and expensive to implement. Incidentally, a so-called blind SLM receiver has been proposed in “SLM and PTS peak-power reduction of OFDM signals without side information”, A. D. S. Jayalath and C. Tellambura, IEEE Transactions on Wireless Communications, vol. 4, issue 5, September 2005, pp 2006-2013. Various improvements to the basic SLM method aimed at achieving reductions in transmitter and receiver complexity and in signalling overhead are described in our co-pending European patent application no. 05256600.7, the entire content of which is incorporated herein by reference.
A further problem with SLM is that, because the set of phase vectors is determined on a random basis, it is unable to reach the ideal (PAPR free) performance, especially when the number of sub-carriers is large. Although the PAPR performance theoretically keeps on improving as the number of phase vectors is increased, for large numbers of phase vectors it is found in practice that the performance improvement becomes saturated at a level below the optimum level.
Another PAPR reduction technique that has also been proposed is a tone reservation (TR) technique. In this technique, some sub-carriers from among the entire set of available sub-carriers are reserved for PAPR reduction. The reserved carriers carry no data. The receiver simply disregards the sub-carriers which carry no data and recovers the data from the remaining sub-carriers. It has also been proposed to apply a gradient algorithm as part of the TR technique. The basic idea of the gradient algorithm comes from clipping. Using the gradient algorithm, signals having an impulse-like characteristic are generated using the sub-carriers that carry no data, and IFFT output signals are clipped using the signals having the impulse-like characteristic. When the generated signals having an impulse-like characteristic are added to the IFFT output signals, data distortion occurs only in some sub-carriers carrying no data and does not occur in the other sub-carriers carrying data.
FIG. 4 shows parts of an OFDM transmitter 80 adapted to carry out the TR technique using a gradient algorithm.
The transmitter 80 comprises a tone allocation unit 82 which has N-L first inputs and L second inputs (in this context “tone” means sub-carrier). Each of the first inputs is connected to a serial-to-parallel converter 84 which operates in the same way as the serial-to-parallel converter 12 of FIG. 1 except that in this case each group of input symbols is made up of N-L input symbols instead of N input symbols in FIG. 1. As in the FIG. 1 transmitter, the group of input symbols may include pilot symbols as well as data symbols. The second inputs of the tone allocation unit 82 are supplied with L peak reduction symbols. Mathematically, these L peak reduction symbols can be represented by L non-zero elements in a peak reduction vector C having N elements in total, the remaining N-L elements of C being zero. Similarly, the group of N-L input symbols applied to the first inputs of the tone allocation unit can be represented by N-L non-zero elements in an input-symbol vector X having N elements in total, the remaining L elements of X being zero. The peak reduction vector C and the input-symbol vector X must lie in disjoint frequency subspaces.
The group of N-L input symbols and the L non-zero values of the peak reduction vector C are applied to different inputs of an IFFT processing unit 88 which carries out N-point IFFT processing on them. The IFFT output signals are applied to a parallel-to-serial converter 90 to produce a series of time-domain digital samples x. These time-domain signals x are applied to a gradient algorithm unit 92 which implements a gradient algorithm. The gradient algorithm is an iterative clipping algorithm using a peak reduction kernel p. p is a function only of the reserved tone locations. p is the IFFT output of the vector P whose value is 1 at the reserved tone locations and 0 elsewhere. When p is circularly shifted, scaled and phase rotated in the time domain, the values of P in the reserved tone locations are changed but the values of X in the other non-reserved locations are unchanged. Accordingly, the input-symbol vector X is not affected by the iterative clipping algorithm. The optimisation is carried out only on the time-domain digital samples. As a result, only one IFFT operation is needed and the complexity is relatively low. Further details of the operation of the gradient algorithm applied to the TR technique can be found, for example, in “Tone reservation method for PAPR reduction scheme”, IEEE 802.16e-03/60rl, Park, Sung-Eun et al. Samsung Elec., IEEE 802.16 Broadband Wireless Access Working Group, Nov. 10, 2003.
The TR technique with gradient algorithm must carry out calculations serially. In order to achieve a good performance gain with a low processing delay, the processing capability of the transmitter must be high. Also, the number of required iterations is generally high (for example around 30 iterations are suggested in the above paper) and this leads to a high processing burden and/or high power consumption.